Coupling devices (referred to as couplers) in general, such as for example Hybrid 3 dB couplers, are essential circuit components which are increasingly being used for high performance applications in such diverse circuits as RF mixers, amplifiers and Modulators. In addition they can be used in a variety of other support functions such as the ones encountered in general RF signal and amplitude Conditioning and error signal retrieval systems.
The expression “hybrid” in connection with couplers means an equal split of power between two (output) ports of the coupler with respect to an input port. Hence a 3 dB coupler is a “hybrid” since:10 log(Powerout/Powerin)=−3 dBPowerout/Powerin=10(−3/10)=0.5
So the output power Powerout of one of the output ports is half (−3 dB) of the input power Powerin, the other half emerges from the other output port. If we consider FIG. 2 (to be explained in greater detail later on) and say that port P1 is the input port, then port P4 is said to be the coupled port and port P2 is said to be the direct port with half the input power being output from each of the output ports. Port P3 is said to be isolated from port P1. Note that the output at the direct port will experience a phase shift dependent on the coupling length while the output at the coupled port will not experience a phase shift (with reference to the input supplied at the input port).
The use of couplers in the 1-5 GHz range though has been at the expense of large area of occupation required for such couplers and fabrication tolerance problems resulting from tight gap dimensioning for 3 dB coupling operation when implemented in PCB technology (PCB=Printed Circuit Board). More precisely, when implementing a coupler in PCB technology, it is necessary to accurately provide a gap between coupling lines of a coupler with the designed dimensions since otherwise the coupler will not perform properly.
To address fabrication issues, narrow-band equivalents that compromise even more the size of the circuit such as branch line couplers have been utilised. Other alternatives such as SMD type (SMD=Surface Mounted Device) hybrid couplers have been used that offer better size ratios but are still quite large for future small size increased functionality systems. Often SMD component type couplers require additional external matching components to optimise their performance in terms of isolation and matching as well as amplitude and phase balance and therefore even further compromise the circuit area. Stated in other words, the provision of externally provided SMD components for matching purposes further increases the entire size of the coupler and requires additional soldering processes for soldering the externally provided SMD components. The increased use of SMD components increases costs and the use of soldering connections compromises the environmental friendliness and reduces the reliability of a manufactured subsystem module since each solder connection represents a potentially source of error.
Stripline technology has also been utilised for the design of high performance couplers but its suffers from the need to accommodate for larger volume/size for a given component inflicting additionally more materials costs.
Low loss performance can also be an issue especially in LNA designs (LNA=Low Noise Amplifier) as well as in high efficiency power amplification and linearisation applications. For such applications a fraction of a dB improvement can be advantageous. Current designs offer typically 0.3 dB loss performance per coupler.
To rectify the above problems and address the performance requirements of future miniaturised circuit subsystems, wideband couplers in terms of isolation, matching and amplitude and phase balance are required that are additionally fabrication tolerance resistant and of much smaller size than its predecessors.
Size can be decreased by using an appropriate integration technology as well as a miniaturisation circuit technique. Multilayer integrated circuits such as multilayer ceramic LTCC/HTCC (LTCC=Low Temperature Cofired Ceramics, HTCC=High Temperature Cofired Ceramics) technologies have been identified as a technology of great miniaturisation potential since three dimensional design flexibility is combined with ceramic materials of high dielectric constant ( ). Loss performance is enabled by the careful choice of materials and circuit geometry as well as topology.
Isolation/matching and amplitude and phase balance performance can be optimised by using a suitable circuit technique or geometry.
FIG. 1 shows an example of a practical multilayer stack-up as known for dense integration in multilayer ceramic technologies such as LTCC/HTCC. As can be seen in FIG. 1, different ground planes, which achieve isolation, separate different integration levels. This high-density integration scenario relies on the use of stripline components, which as stated above suffers from an increase in area/volume for a given component inflicting additionally more material costs.
Stated in other words, Monolithic integration of passive components into multilayer passive substrates is highly useful for addressing the size, cost, and performance trade-offs that dominate much of the design efforts in the mobile telecommunication industry. Low Temperature Cofired Ceramic (LTCC) technology is an important example of an available multilayer substrate. FIG. 1 shows in rough outline an example of a practical multilayer stack-up in LTCC using two different ceramic thicknesses. The top substrate layer is utilised for bias and wirebound MCIC circuitry, with the bottom layer used for soldering packaged components (e.g. using ball grid array BGA). The two middle layers are used for controlled impedance transmission lines and other passive components such as parallel plate capacitors, inductors, couplers, baluns and power dividers. It can be seen that for adequate isolation between circuitry of different design layers the passive elements need to be implemented as stripline components with different design layers separated by ground planes.
The implementation of stripline couplers carries a significant disadvantage in requiring a much larger thickness of substrate as compared to its microstrip counterpart to achieve similar performance for the same geometry. Hence when optimising for cost by reducing the number of layers used, the performance of stripline couplers will suffer.
FIG. 2 shows an equivalent circuit diagram of a conventionally known coupler. Basically, a coupling device consists of a pair of coupled lines 3a, 3b. Each line has two ports for inputting/outputting electrical and/or electromagnetic signals to be coupled. Thus, as shown in FIG. 2, the line 3a has ports P1, P2, while the line 3b has ports P3, P4. Each port P1 through P4 is terminated with a termination impedance Z0. In a 50 Ohms system, the value of Z0 is set to 50 Ohms. The lines 3a, 3b have equal length which is expressed in terms of the wavelength for which the coupler is designed. The parameter le° denotes the electric length of the coupler which is measured in degrees (°). For example, for the coupler shown in FIG. 2 the length is assumed to be λ/4, with λ/4 corresponding to the center frequency of operation for which the coupler is designed. Thus, in such a case, a signal fed to the coupler at port P1 and used as a reference (indicated by “0°”) is coupled to the port P4 (coupled port) with its phase unaltered. Port P3 is isolated from port P1, which means that no power reaches port P3 from port P1. The signal at port P2 (the direct port) is shifted with reference to the signal input at port P1 as indicated by +90°. Note that in case of a 3 dB coupler as an example, the power input at port P1 is split between ports P2 (direct port) and P4 (coupled port). Nevertheless, other line lengths such as λ/2, or odd multiples of λ/4 such as 3λ/4 are possible. Also, the lines could have different lengths, while in such a case only the length of the lines over which coupling takes place represents an effective coupling length (electric length le in [°] of the coupler). The coupler, i.e. the coupling lines, may be described in terms of the even and odd propagation modes of electromagnetic waves travelling there through and their respective characteristic impedances Zoo, Zoe and phase velocities υoe and υoo and the electric length le of the coupling lines.
In 3 dB coupling in a 50 Ohms system, one needs to design the lines to have impedance values Zoo and Zoe of 20.7 and 120.7 Ohms respectively. The above arrangement though assumes equal phase velocities for the even and the odd modes i.e. υoe=υoo. This assumption holds for homogeneous couplers such as stripline couplers.
Zoe is primarily effected by the thickness of the substrate and transmission line widths. Often, in practical implementations, the substrate thickness is less then that required for achieving the correct Zoe. This may be due to size, cost or reliability considerations, or a combination of all. The reduced Zoe impacts adversely on the amplitude and phase balance of the coupler, as well as on the matching and isolation.
Reduction in Zoe can be dealt with in two ways, either we increase the substrate thickness incurring significant material costs and increasing the volume of the component; or reducing the transmission line width, which is limited by manufacturing requirements and tolerance limitations. Reducing the transmission line width has an adverse effect on Zoo, which imposes a limit on how much we can ultimately reduce the width and still be able to meet the Zoo requirement.
The present invention to be described herein below is described with reference to stripline couplers. Nevertheless, the proposed structural modification according to the present invention is also applicable to microstrip couplers. Also, it is not essential for the present invention whether broadside coupled or edge coupled couplers are concerned. However, in order to describe the present invention, a focus in the description is laid on broadside coupled stripline couplers, without imposing any limitation on the invention.
FIG. 12 shows in a rough outline the basic difference between a stripline and microstrip arrangement, respectively. The left hand portion of FIG. 12 shows a stripline arrangement, while the right hand portion shows a microstrip arrangement (both edge coupled as the conductive layers are placed in the same layer with the edges facing each other). It is an important property of any lossless coupled transmission lines (coupling lines) placed in a uniform dielectric substrate (homogeneous substrate and/or symmetrical) that it supports a pure TEM mode of propagation. A common example of these types of lines is STRIPLINE, as shown in FIG. 12, left portion. However if a transmission line is placed in an inhomogenous (and/or non-symmetric) dielectric substrate it can no longer support fully-TEM propagation because the electromagnetic wave-now propagates mostly within the substrate, but some of the wave is now able to propagate in air also. The most common example of this is MICROSTRIP also shown in FIG. 12, right portion. Stripline couplers are encased in a homogenous substrate where the electromagnetic fields of the coupler are confined within the substrate by the two ground planes. While for a microstrip line its electromagnetic propagation takes place mainly within the substrate (in fact most of the power propagates within the substrate), but some of the power propagates outside the substrate which is usually air.
FIG. 3 shows basic structural arrangements in cross section of broadside coupled structures. FIG. 3 shows the typical structures utilised in the design of Hybrid-Couplers in Multilayer ceramic technology. The Broadside Coupled structures are a very useful design structure that can adjust the amount of coupling by offsetting the two coupled-transmission lines. FIG. 3 comprises FIGS. 3a, b, c, and d illustrating (FIG. 3a) a broadside coupled stripline (without offset between coupling lines), (FIG. 3b) an offset-broadside coupled stripline, (FIG. 3c) a broadside coupled microstrip (without offset between coupling lines), (FIG. 3b) an offset-broadside coupled microstrip.
Thus, as shown in FIG. 3c and d, a respective coupling device, comprises a substrate 1, a first conductive layer 2 covering a first surface of said substrate 1, at least two electromagnetically coupled lines 3a, 3b being provided opposite to said first surface and being covered by at least one cover layer 4, 5. Additionally, as shown in FIG. 3a and b, said at least one cover layer 4, 5 is covered by a second conductive layer 2′. Said at least two lines 3a, 3b are arranged at different distances from said first surface of said substrate 1, wherein the difference between the distances in which said at least two lines 3a, 3b are arranged from said first surface of said substrate 1 is determined by a thickness of a first cover layer 4 covering a first line 3b of said at least two lines. As shown, the first line 3b and a second line 3a of said at least two lines are arranged such that they at least partly overlap each other (FIG. 3b and d), the amount of overlap adjusting the degree of electromagnetic coupling between said at least two lines.
A second cover layer 5 is arranged to cover at least a second line 3a of said at least two lines. Of course, said at least one cover layer 4, 5 can be of the same material as said substrate 1, which is made of a dielectric material of a relative dielectric permittivity εr. Said conductive layers 2, 2′ are connectable to ground potential.
FIG. 4 shows a specific comparative example for comparison with the present invention (still to be described later in this document). The example of FIG. 4 is based on a broadside coupled stripline coupler as previously shown in FIG. 3a above. In detail, FIG. 4 shows a perfect broadside-coupled stripline coupler. The results shown derive from momentum-based simulations (2.5-D EM simulator). The coupler is designed to exhibit Zoe=120.7 Ohms and Zoo=20.7 Ohms with Ve=Vo, at a central frequency of 1.8 GHz. However, to achieve this response an LTCC substrate of thickness 2.3 mm, with εr=7.8 (and layer 4 thickness of 0.094 mm) was required. The coupler according to FIG. 4 is 15 mm in length to achieve the required central frequency. However, one can reduce this length by meandering the coupler as shown in FIG. 5, which shows a further comparative example. Meandering introduces structural discontinuities which degrade the performance by introducing asymmetry for the normally symmetrical normal modes of propagation. This manifests itself as an inequality in the normal-mode phase velocities, Ve≠Vo. This accounts for the reduction in performance observed in FIG. 5.
If the substrate thickness is reduced by more than half from 2.3 mm to 1.1 mm, one can observe a further reduction in the performance of the stripline broadside-coupled coupler. This is due to the degradation of Zoe, which is therefore to be compensated for.
Note also, that as the production technology for such devices the multilayer integrated circuit technology which is assumed to be well known to those skilled in the art may be used so that a detailed description of the method for production of such devices is considered to be dispensable.
To the best of our knowledge there have not been suggested any techniques that compensates the degradation of Zoe.